The present invention concerns radio receivers used for digital communication. It has particular, although not exclusive, applicability to matched filters used in such receivers.
Digital symbol transmission over the narrow-band channels allocated, for instance, to cellular-telephone traffic imposes stringent filter requirements on the transmitters and receivers used for such communication. The stringency is a result of the need to extract symbols from the transmitted signals without suffering errors that could result from intersymbol interference.
Such requirements can be understood by reference to FIG. 1, which depicts a typical transmitter and receiver employed for digital communication. The example system is of a type that might be used to implement the IS-54 standard proposed for North American cellular-telephone communication. A handset microphone 10, for instance, produces an analog signal that front-end-processing circuitry 12 converts into digital form, encoding it in ways not relevant to the present invention. The result is a bitstream that a .pi./4-shifted differential quadrature phase-shift-keying ("DQPSK") modulator 14 receives in two-bit pairs. Modulator 14 generates a complex-number output dictated by that bit pair in accordance with the DQPSK scheme. Specifically, the complex output is one of eight points equally spaced about the unit circle in the complex plane, and each of the four possible input bit pairs represents a different angle shift from the previous modulator output, the four possible shifts being plus or minus .pi./4 and plus or minus 3.pi./4.
Analog modulators 16 and 17 modulate in-phase and quadrature components, respectively, of a carrier that a source 18 generates, and respective bandpass filters 19 and 20 pass the desired sum or difference components of the resultant signal. If the analog version of the raw modulator-14 output were applied directly to the amplitude modulators 16 and 17, the abrupt phase changes would cause the spectrum of the resultant signal to be much wider than is allowed for single-channel cellular-telephone communication. Digital filters 21 and 22 therefore subject the two DQPSK-modulator outputs to low-pass filtering before the resultant signals are converted to analog form in digital-to-analog converters 23 and 24 and used to modulate respective carrier components. Accordingly, when an adder 25 combines the two thus-modulated carrier components, the bandwidth of the resultant signal is within limits. An amplifier 28 increases that signal's power for transmission over an antenna 30.
An unfortunate but unavoidable result of the filtering that filters 21 and 22 perform is a "smearing" of the sharp pulse outputs of the modulator 14; the filter's response to each such pulse is significant for many symbol periods. Generally, therefore, the output signal at any given time contains contributions from the responses to several input symbols: there is inter-symbol interference.
At the receiver end, a receiving antenna 32 applies its output to a receiver front end 34, whose intermediate-frequency output is mixed in (typically, but not necessarily, analog) mixers 36 and 38 with respective intermediate-frequency signals, produced by a source 40, that are equal in frequency but 90.degree. out of phase with each other; i.e., the intermediate-frequency signal is mixed with a complex sinusoid. Low-pass filters 42 and 44 extract the baseband results of the mixing process, respectively producing in-phase and quadrature components.
We interrupt the receiver description at this point to observe that, although the preceding receiver components are, of course, typical, a receiver, as we use that term here, may be a device used in the transmitter itself to extract symbols from the transmitted signal for certain feedback purposes such as predistortion adjustment. For present purposes, therefore, a receiver does not necessarily include an antenna and related front-end components, although most receivers to which the present invention is applicable will include them.
The outputs of filters 42 and 44 are analog signals, which respective analog-to-digital converters 46 and 48 convert to digital form to produce respective data streams that so-called matched filters 50 and 52 filter digitally. A decision device 54 then determines, from the levels in the matched-filter outputs, what symbols the DQPSK modulator 14 initially produced, and it thereby determines what the modulator's inputs were. The decision device typically would also reverse the front-end processing performed by circuit 12.
Although the segregation between digital and analog processing and between in-phase and quadrature sections described above is typical, the conversion to the digital domain can in principle be performed earlier in the processing chain, before the processing is bifurcated into parallel in-phase and quadrature sections. For instance, a single analog-to-digital converter could convert receiver 34's intermediate-frequency output directly to digital form, and the mixing with the complex sinusoid could be performed in the digital domain. In that case, there would ordinarily be no need for elements corresponding to low-pass filters 42 and 44 separate from the matched filters 50 and 52, which themselves perform a low-pass function.
As was mentioned above, inter-symbol interference results from the pulse-shaping process that filters 21 and 22 perform. Although the resultant "smearing" of the incoming pulses is unavoidable, judicious selection of filter parameters enables the receiver to avoid the confusion that this interference would otherwise cause. Specifically, filters 50 and 52 form "Nyquist filters" with pulse-shaping filters 21 and 22, respectively, in the transmitter. A Nyquist filter is one whose impulse response has zero crossings at delays equal to all non-zero numbers of symbol times. There is thus a point in each symbol time at which the responses of the filter to all of the input pulses except the current one are zero, and if the outputs of the matched filters 50 and 52 are "sampled" only at those times, the effects of inter-symbol interference can be avoided.
(For the sake of convenience, we refer to filters 50 and 52 as "matched" filters because as a practical matter they conventionally are the same as the pulse-shaping filters 21 and 22 and thus match those filters' impulse responses. In principle, however, they need not be the same, so we use matched filter here to mean filters used in producing the Nyquist response, regardless of whether their characteristics are the same as those of the transmitter's pulse-shaping filters.)
The Nyquist-filter approach, which is the one that typically is employed to extract the symbols despite intersymbol interference, naturally depends on accurately approximating ideal Nyquist filters. Among other things, this means that the relative timing of the digitally implemented transmitter and receiver filters must be proper. The transmitter timing is based on a transmitter clock 56, while that of the receiver is based on a separate, independent receiver clock 58. The relative timing between the filters therefore will tend to drift unless appropriate countermeasures are taken.
To prevent such drift receivers of this type conventionally employ a timing-recovery circuit 60 to measure the timing error so that appropriate compensation can be performed. One type of timing-recovery circuit receives the output of one of the matched filters 50 and 52 and digitally correlates it with a complex sinusoid whose frequency is half that of the symbol frequency. This sinusoid is produced by a source 62 timed by the receiver clock 58. For the IS-54 scheme, in which the symbol frequency is 24.3 kHz, the frequency of the complex sinusoid is 12.15 kHz. The source 62 applies its output values to a digital multiplier 64, which multiplies these complex values by the real-value output of, say, matched filter 52 and applies the result to a digital low-pass filter 66.
The output of this low-pass filter 66 is a complex number representing the amplitude a d phase of the 12.15-kHz component present in the output of filter 52. Because of certain aspects of the higher-level encoding performed in the front-end processor 12, this component is ordinarily significant in the resultant signal, and, averaged over time, it indicates the relative timing between the sample times and the symbol times in the filter output and thus whether the phase of the clock is that required to minimize intersymbol interference. Accordingly, an averaging circuit 68 determines the average value of this component, which is then used for timing adjustment.
As those skilled in the art will recognize, various of the digital elements, such as the filters 46 and 48, the multiplier 64, etc., are typically implemented in a common digital signal-processing circuit. Therefore, most of the digital elements do not individually add significantly to the size, cost, or power requirements of the receiver. But a significant addition can be caused by the use of, say, a variable-frequency oscillator for adjustment of the clock phase in response to the recovered timing offset. This is particularly disadvantageous in a cellular-telephone receiver, in which space and power consumption are at a premium.
Rather than employ such a circuit, therefore, a typical receiver employs a fixed clock 58 whose output is not directly used for timing but instead is applied to a sample-pulse generator 70 that includes a tapped delay line so as to impose a variable delay between clock 58 and the circuits that require the timing signals. The tap output used as the sample pulse is selected in accordance with the output of averaging circuit 68. That is, if the phase is proper, the tap remains the same, but it is advanced or retarded if the phase is improper. As a result, the sample-pulse generator adjusts for the clock phase difference so as to ensure that the matched filters 50 and 52 form respective Nyquist filters with the pulse-shaping filters 22 and 23.
The use of a variable-delay-line type of pulse generator 70 yields space advantages over, say, a variable-frequency oscillator, but it still requires extra space. Moreover, the timing resolution that it can achieve is ordinarily not as good as that of a variable-frequency oscillator; it can achieve only discrete values of delay, and the number of delay values is limited in a reasonable-sized tapped delay circuit.